Apparatus and method for embedding current measurement and ringing suppression in multichip modules

ABSTRACT

The current disclosure relates to the design of an apparatus for enhancing the operation and reliability of high-power multi-chip modules, which are used in the design and implementation of power electronics converters. This apparatus is especially useful for modules containing recently commercialized, high-performance wide band-gap semiconductors such as Silicon Carbide (SiC), which commonly emit undesirable high-frequency ringing and oscillation in the “Near-RF” spectral band between 1-30 MHz. The disclosed apparatus provides near-complete elimination of this high frequency spectral content, while leaving the desired frequency range (1-100 kHz) of the module unaffected. In addition to the suppression of this undesirable high-frequency content, the disclosed apparatus also provides for accurate, galvanically-isolated, high-bandwidth, real-time current measurement, which is essential for some types of power electronics converters. The apparatus disclosed here provides ringing suppression and current measurement in simple circuit topology that can be implemented compactly inside the geometry of a multi-chip power module.

BACKGROUND Field

The present disclosure relates to a current transformer for powerelectronics applications, specifically a current transformer that alsoprovides suppression of undesired signal content, in addition to theprincipal role of the current transformer, which is the measurement ofcurrent.

With the commercialization of wide band-gap (WBG) semiconductors, a newpromising horizon has been opened to power electronics applicationdesigners. For the full potential of this technology to be exploited forhigh-power applications, multi-chip power modules (MCPM's) are required.MCPM's bring provide several advantages to high-power applications,including compactness, thermal performance, mechanical stability, andincreased operational life. Efforts to optimize MCPM's for wide band-gapsemiconductors have made significant progress over the last severalyears; however, some challenges remain. For example, the very fast di/dtand dv/dt rising edges, introduced by Silicon Carbide (SiC) MOSFET's,can easily excite resonances in parasitic elements, causing ringing andovershoot during the switching transients, and consequently impeding thedesired application functionality. Even for an optimized structure suchas an MCPM, in which parasitic inductances of traces have been minimizedto the extent possible, this challenge usually remains a concern [1].Therefore, additional techniques to mitigate this behavior and improvethe transient response of WBG-based MCPM's are needed.

On the other hand, in many power converter topologies, high-bandwidth,real-time current measurement is a requirement for control. Severaloptions are available to engineers for this purpose [1][3]. Resistiveshunts [4][5] offer high bandwidth but no galvanic isolation;traditional current transformers (CT) [6][7] provide galvanic isolationbut require a mechanical choke-point in the power bus which increasesloop inductance (which degrades the transient behavior of WBG circuits)and exacerbates undesirable signal content; and hall-effect sensors [8][9] are only suitable for measuring currents with modest bandwidth(kHz).

BRIEF SUMMARY

Accordingly, the present disclosure is directed to apparatus and methodfor embedding current measurement and ringing suppression in multichipmodules that obviates one or more of the problems due to limitations anddisadvantages of the related art.

In accordance with the purpose(s) of this disclosure, as embodied andbroadly described herein, this invention, in one aspect, relates to acurrent measurement and ringing suppression device for use in amulti-chip power module which includes a magnetic core currenttransformer having an insertion impedance Zin and turns ratio N, aburden resistor having resistance R_(B) and a filter, the filtercomprising a capacitance C_(P), an impedance L_(P) and a resistanceR_(P); wherein C_(P), L_(P) and R_(P) are selected such that Zin remainsabove a minimum effective value across a known frequency range; whereincurrent measurement is taken across the burden resistor.

In another aspect, the invention relates to a multichip power modulewhich includes a plurality of multi-chip power module terminals; amagnetic-core-based current transformer having an insertion impedanceZin and turns ratio N, a burden resister having resistance R_(B) and afilter, the filter comprising a capacitance C_(P), an inductance L_(P)and a resistance R_(P); wherein C_(P) L_(P) and R_(P) are selected suchthat Zin remains above a minimum effective value across a knownfrequency range; wherein current measurement is taken across the burdenresistor; wherein the magnetic core current transformer is positionedaround a portion of at least one of the multi-chip power moduleterminals.

In yet another aspect, the invention relates to a method of measuringcurrent and snubbing in a multi-chip power module, the multi-chip powermodule comprising a magnetic core current transformer having aninsertion impedance Zin and turns ratio N, a burden resister havingresistance R_(B) and a filter, the filter comprising a capacitanceC_(P), an inductance L_(P) and a resistance R_(P); wherein C_(P), L_(P)and R_(P) are selected such that |Zin|≈N²·R_(B), the method comprisingattaching the current transformer to the system for which measurement isdesired; and measuring the voltage across the burden resistor RB; andmathematically scaling the voltage across the burden resistor RB torepresent the predicted current value in the primary circuit.

Additional advantages of the invention will be set forth in part in thedescription which follows, and in part will be obvious from thedescription, or may be learned by practice of the invention. Theadvantages of the invention will be realized and attained by means ofthe elements and combinations particularly pointed out in the appendedclaims. It is to be understood that both the foregoing generaldescription and the following detailed description are exemplary andexplanatory only and are not restrictive of the invention, as claimed.

An advantage of the present invention is to provide apparatus and methodfor embedding current measurement and ringing suppression in multichipmodules.

Further embodiments, features, and advantages of the apparatus andmethod for embedding current measurement and ringing suppression inmultichip modules as well as the structure and operation of the variousembodiments of the Apparatus and method for embedding currentmeasurement and ringing suppression in multichip modules, are describedin detail below with reference to the accompanying drawings.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory onlyand are not restrictive of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying figures, which are incorporated herein and form part ofthe specification, illustrate apparatus and method for embedding currentmeasurement and ringing suppression in multichip modules. Together withthe description, the figures further explain the principles of theapparatus and method for embedding current measurement and ringingsuppression in multichip modules described herein and thereby enable aperson skilled in the pertinent art to make and use the Apparatus andmethod for embedding current measurement and ringing suppression inmultichip modules.

FIG. 1 illustrates a CT-snubber according to principles of the presentdisclosure in a multi-chip power module (MCPM).

FIG. 2 is a schematic of an exemplary embodiment of a CT-snubber circuitaccording to principles described herein.

FIG. 3 is a graph illustrating exemplary insertion impedance ofCT-snubber in the frequency domain and associated bands of influence.

FIGS. 4(a) and 4(b) illustrate current measurement effectiveness of aCT-Snubber according to principles of the present disclosure. In thisfigure, the signal from a reference instrumentation shunt (dashed trace)is compared to the current signal predicted by the CT-snubber (solidtrace).

FIG. 5 shows a test prototype CT-Snubber according to principles of thepresent disclosure.

FIG. 6 shows an empirical impedance and a simulated impedance profile ofan ideal parallel RLC filter, such as that incorporated into theexemplary CT-snubber embodiment described herein.

FIG. 7A shows a simulated plot of the frequency-dependent insertionimpedance (Zin) of the CT-snubber when the filter capacitance (C_(P)) isvaried across a range from 1 pF to 100 pF.

FIG. 7B shows a simulated plot of the frequency-dependent insertionimpedance (Zin) of the CT-snubber when the filter inductance (L_(P)) isvaried across a range from 30 pH to 90 μH.

FIG. 7C shows a simulated plot of the frequency-dependent insertionimpedance (Zin) of the CT-snubber when the filter resistance (R_(P)) isvaried across a range from 300Ω to 400Ω.

FIG. 8 is a schematic of a clamped-inductive load (CIL) test stand, withCT-snubber implementation.

FIG. 9 shows the physical realization of a CIL test stand for testing aprototype CT-snubber according to principles of the present disclosure.

FIG. 10 shows experimental waveforms at turn-off, with and without theCT-snubber in place: (a) current waveforms, (b) voltage waveforms.

FIG. 11 shows experimental waveforms at turn-on, with and without theCT-snubber in place: (a) current waveforms, (b) voltage waveforms.

FIG. 12 illustrates the current measurement capabilities of the presentinvention by comparing the signal from a reference instrumentation shunt(dashed trace) to the current signal predicted by the CT-snubber (solidtrace).

DETAILED DESCRIPTION

Reference will now be made in detail to embodiments of the apparatus andmethod for embedding current measurement and ringing suppression inmultichip modules with reference to the accompanying figures, in whichlike reference numerals indicate like elements.

According to principles of the present disclosure, a current transformer(CT) can be fitted around an internal terminal within a multi-chip powermodule (MCPM). According to novel principles described herein,high-bandwidth current measurement can be performed without the need forintroducing an undesirable geometric choke-point in the externalbussing.

An illustration of an exemplary CT-snubber 100 embedded in the MCPM 102is provided as FIG. 1. As can be seen in the illustration, the CT isembedded around the terminals 103 of the MCPM, taking advantage of anexisting, necessary constriction in the system where current flowsthrough the terminals, rather than establishing another choke-pointspecifically for current measurement, as in conventional designs (e.g.to enter the aperture of the CT).

The greater opportunity, moreover, arises when recognizing that thefrequency response of a traditional CT can be described as a bandpassfilter bounded by minimum and maximum cut-off frequencies. The bandpassis where the content of interest normally resides for accurate CT-basedcurrent measurement (the “measurement band”). However, by carefulselection of current transformer (CT) properties and using a fewadditional components, this pass band can also be accompanied by a“suppression band”, effectively utilizing the CT insertion impedance tosuppress high-frequency (usually MHz range) spectral content. In thisway, the undesirable high-frequency spectral content is suppressed,while still providing an accurate current measurement of the desired,lower-frequency content. This provides a similar damping effect to thatoffered by a traditional bus snubber, but with minimal impact for thedesired converter behavior, and while simultaneously enabling agalvanically isolated, high-bandwidth current measurement. This solutioncan also be embedded within the MCPM packaging with minimal effect onthe dimensions of the device. Moreover, the CT-Snubber will generallyprovide results similar to a traditional CT measurement in themeasurement band. That is, frequency dependence of the CT can be used todiscriminate between desired and undesired frequency components in thesignal.

In a traditional CT design, there are only two degrees of freedomavailable to the designer to shape the bandpass of the sensor as well asthe insertion impedance of the CT: the burden resistor value and thesecondary turns ratio. A “CT-snubber” according to principles of thepresent disclosure contains additional degrees of freedom to providefurther flexibility for shaping the frequency response of the sensorgain and the insertion impedance. This additional flexibility allowsfor: 1) measuring the current within a defined “measurement band” withsufficient accuracy, as will be described further herein; and 2)snubbing out or reducing undesirable high-frequency spectral content ina defined “suppression band”. As rule of thumb, in many powerelectronics applications, the undesired high-frequency spectral contentcan usually be identified as that starting one order of magnitude higherthan the switching frequency. This can be leveraged as a guideline todesign any effective snubber. Specifically, an effective snubber isexpected to present high impedance above a certain cut-off frequency toimpede high frequency currents, while at the same time showing very lowimpedance below that cut-off frequency in order to avoid any negativeside effects on the normal operation of the system.

At frequencies that fall into the measurement band, measurement behavioris expected from the CT-snubber, meaning that it should exhibit a lowinsertion impedance. This impedance should not fall below a certainpoint or the expected current measurement functionality may becompromised. At frequencies within the suppression band, the CT-snubbershould exhibit high insertion impedance to suppress the undesiredspectral content associated with parasitic-induced ringing. Therefore,it should be understood that each of these frequency regions should beadjusted properly in order to achieve an effective solution for anMCPM-embedded current measurement with ringing suppression capability.

An equivalent circuit of the proposed CT-snubber 100 is depicted in FIG.2. The current transformer 204 itself is represented by an idealtransformer, accompanied by a burden resistor 208, RB, and a parallelRLC filter 212 in the secondary to realize separation of the frequencyresponse into a measurement band and a suppression band. The proposedCT-snubber 100 includes a magnetic coupler to provide a definedelectrical relationship between the application (or “primary”) sidecircuit and the measurement (or “secondary”) side circuit. The primarycoil has a first number of windings “turns” Np and the secondary coilhas a second number of winding “turns” Ns. As is common in the art, theprimary coil is comprised of a single turn, represented by a conductorin the application circuit which passes through the CT aperture. Thesecondary coil generally has a larger number of turns; a typical rangeof Ns for the present invention is in the range 10-100.

In general, the insertion impedance of the CT-snubber can be derived as:

$\begin{matrix}{Z_{in} = {N^{2} \cdot ( {R_{B} + \frac{R_{P}}{1 + {j( {{\omega \; C_{P}R_{P}} - \frac{R_{P}}{\omega \; L_{P}}} )}}} )}} & (1)\end{matrix}$

Where N is the current transformer turns ratio (i.e. NP/NS); R_(B) isthe burden resistor; and C_(P), L_(P), and R_(P) are the components ofthe parallel RLC “filter” portion of the CT-snubber shown in FIG. 2.

In this context, according to principles of the present disclosure, theRLC filter 212 causes the CT insertion impedance Z_(in) to spike atfrequencies greater than the frequencies at which current measurement isperformed, allowing the insertion impedance Z_(in) to be separated intofour different regions (frequency domains), as shown in FIG. 3. Thefirst region is the DC band in which the insertion impedance of CT isalmost zero due to the fact that a transformer is shorted out for verylow frequencies. In this DC Band region 316, the CT-snubber is neitherable nor expected to carry out any measurement:

|Z _(in,DC)|≈0  (2)

The second region, known as the measurement band 320, is where accuratemeasurement performance is expected. One of the primary design goals fora traditional CT is attainment of a flat gain profile for the specifiedmeasurement bandwidth. Generally, such design goal for a traditional CTmitigates measurement error, which would otherwise accrue due tofrequency-dependent CT gain. It should be noted that all CT's havefrequency-dependent gain, since they exhibit gain roll-off both atlow-frequency due to “droop” and at high-frequency due to shuntcapacitance and magnetic circuit limits.

For the case of a CT with embedded suppression capability as describedherein, the current measurement may be extracted from the CT-snubber bysampling the voltage across the burden resistor (R_(B)), just as in atraditional CT. Therefore, the RLC filter which appears in series withthis element does not directly introduce measurement error in the senseof causing the measurement result to diverge from the physical behaviorof the circuit. Provided that the transformer remains linear and theburden resistor (R_(B)) is reasonably modeled by an ideal resistanceacross the entire measurement band, the output (V_(SENSE)) will be asufficiently accurate, scaled representation of the primary-sidecurrent; irrespective of whether any influence is exerted over theprimary-side circuit by the CT-snubber (such as ringing suppression).

The RLC filter can influence the behavior of the primary-side currentand cause the circuit to behave differently than it would without theCT-snubber in place. That is to say, the RLC filter provides a known“insertion impedance” which can restrict the flow of current within theprimary portion of the CT-snubber in a manner which is frequencydependent. The frequency-dependent insertion impedance profile of theCT-snubber can be tailored by a designer to suppress known frequencyranges which contain undesirable content, such as the 1-30 MHz range,which is known to contain significant spurious emissions in applicationsbased on WBG-based power electronics [12]. Within the measurement band,such an influence would be undesirable in part because it could resultin significant dissipation within the CT-snubber. Assuming that the RLCfilter is designed to exert minimal influence within the measurementband, the CT-snubber insertion impedance in this region can be definedsolely by RB and the turns ratio:

|Z _(in,m) |≈N ² ·R _(g)  (3)

In the suppression band, it is intended for the CT-snubber to change itsrole to suppress undesirable spectral content. This goal is realized byincorporating a parallel RLC circuit in series with the burden resistor108 as shown in FIG. 2. The RLC circuit is responsible for demarking thestart and end frequencies for the high insertion impedance required forsuppressing undesirable high-frequency content. As already mentioned,however, the impedance introduced by the parallel RLC should be verysmall within the measurement band to avoid impact on the intendedoperation of the circuit. A parallel RL circuit, instead of the parallelRLC circuit, can also be employed for this purpose. However, a practicalimplementation of any inductor is accompanied by some inter-windingcapacitance, which can be modeled as equivalent parallel capacitance(EPC). At high frequencies, where the impedance of the EPC becomescomparable to the other components of a parallel RL circuit, theinfluence of this parasitic element cannot be neglected. Therefore, anypractical implementation of a parallel RL circuit effectively turns intoa parallel RLC circuit beyond a certain frequency. In this analysis, thecapacitance in the depicted parallel RLC circuit represents thecombination of inductor EPC and any additional capacitors that designermay need to employ. In the suppression band 328, the insertion impedanceis dominated by the parallel RLC circuit, given as

$\begin{matrix}{{Z_{{in},z}} \approx {N^{2} \cdot \frac{R_{P}}{\sqrt{( {1 + ( {{\omega \; C_{P}R_{P}} - \frac{R_{P}}{\omega \; L_{P}}} )^{2}} }}}} & (4)\end{matrix}$

Also, as shown in FIG. 3, another region can also be identified in theinsertion impedance profile of CT-snubber: the “transition band” 324. Inthe transmission band, the RLC filter 212 begins to exert some influenceover the primary side of the circuit, but high insertion impedance isnot yet achieved. The presence of this band could cause difficulty ifthe application has intended frequency content which is not wellseparated from the content desired for suppression. Fortunately, mostpower electronics applications involve significant separation betweenthe desired content (at the switching frequency) and the undesiredcontent (parasitic ringing). The transition band 324 can be described bythe regime between two cut-off frequencies, ωc1 and ωc2, which exist atthe upper and lower ends of this band 324. In general, the boundaries ofthis transition band will be defined on an application-specific basis.However, it might be reasonable to specify these boundaries in terms ofthe insertion impedance of the measurement band.

For example, ωc1 and ωc2 could be defined as the frequencies at whichthe insertion impedance reaches a factor of 1.05 and 10 times, themeasurement-band insertion impedance, respectively:

ω_(c1)=ω|_(|Z) _(in) _(|=1.05×|Z) _(in,m) _(|)  (5)

ω_(c2)=ω|_(|Z) _(in) _(|=10×|Z) _(in,m) _(|)  (6)

The provided insertion impedance range is merely exemplary. In certainapplications, a designer may determine that a different insertionimpedance magnitude and/or is appropriate.

Current measurement effectiveness of a CT-Snubber according toprinciples of the present disclosure are illustrated in FIGS. 4(a) and4(b). FIG. 4(a) is an example of rapidly decreasing current(semiconductor turn-off), and FIG. 4(b) is an example of rapidlyincreasing current (semiconductor turn-on). These figures demonstrate acomparison between the output of a traditional high-accuracy currentmeasurement device (represented as “REF Measurement”) and the currentmeasurement output of the CT-Snubber. The good agreement observedbetween these two waveforms in this figure verifies that the CT-Snubberis able to accurately predict the current in the primary circuit, aswould be expected for a traditional CT. It should be noted that theCT-snubber is able to accurately predict both the low-frequency (largeamplitude) change as well as the high-frequency (oscillation) behaviorof the primary circuit current. This example verifies that theCT-snubber has sufficient bandwidth to measure primary circuit currentacross a wide range of frequencies with high fidelity.

A set of empirical procedures was carried out to confirm the projectedbehavior of a CT-snubber device according to principles describedherein. This empirical analysis consisted of two distinct operations.First, an exemplary CT-snubber 500, shown in FIG. 5, was characterizedin the frequency domain to evaluate that impedance in the measurementband could be distinguished from impedance in the suppression band.Second, the exemplary CT-snubber was integrated into a high-powerclamped-inductive load (CIL) test stand and its transient performancewas evaluated during double-pulse testing of a high-performance siliconcarbide (SiC) power module. The CT-snubber 500 was evaluated both forcurrent measurement and for suppressing high-frequency ringing.

An exemplary CT-snubber module according to principles described hereinwas characterized in the frequency domain using a 120-MHz precisionimpedance analyzer, which is considered well-suited for determining thefrequency-dependent profile of arbitrary linear circuit networks,including the effects of unknown parasitic elements.

The CT-snubber hardware prototype is shown in FIG. 5. It should be notedthat an earlier CT-snubber prototype (not shown) included both a burdenresistor, as well as a parallel RLC filter network, as describedpreviously in this paper. However, after the characterization processoutlined here, it was determined that the inter-winding parasiticcapacitance of the inductor within the CT-snubber was contributingsignificant capacitance to the parallel RLC network. This parasiticcapacitance was found to be approximately an order of magnitude greaterthan the designed capacitance value of the filter. Therefore, theinductor was re-wound more sparsely with smaller gauge wire in order tominimize the turn-to-turn capacitance; and the discrete capacitor wasremoved from the CT-snubber.

The CT-snubber prototype shown in FIG. 5 was measured across a frequencyrange of 500 Hz to 100 MHz. The resulting empirical impedance profile isshown in FIG. 6, overlaid with the impedance profile of an idealparallel RLC network. It should be noted that this measurementrepresents the impedance of only a portion of the CT-snubber: theparallel RLC filter combined in series with the burden resistor (RB).The ferrite-based structure used to magnetically couple the CT-snubberto the main power circuit is not characterized in this step. Measurementof the AC-coupling interface presents some challenges, due to thereduced impedance magnitude reflected through the coupling, as well asthe frequency dependent characteristics of the magnetic core materialused in the coupling mechanism. However, to first order, the insertionimpedance of the CT-snubber within the main power circuit can be viewedas the impedance profile shown in FIG. 6, with the magnitude divided byfactor of 100 (for a 10-turn secondary winding on the coupler). That is,FIG. 6 shows empirical impedance curves for CT-snubber prototype andparallel RLC model overlay with R_(P)=318Ω, L_(P)=47.8 μH, C_(P)=50 pF.

It should also be noted that the magnetic material employed in thedesign of the CT-snubber is known to play a significant role indetermining the snubbing efficacy of the system. In addition to thefrequency-dependent impedance introduced by the RLC filter (212), themagnetic coupler (204) also provides a measure of frequency-dependentimpedance to the primary-side circuit. This influence is difficult toutilize as a design variable, since unlike the RLC filter, the frequencydependence of the coupler cannot be predicted by linear circuit analysisbut instead is convolved with the material properties of the magneticcore. Nevertheless, the magnetic coupler can significantly enhance thesnubbing effectiveness of the system, especially in the Megahertzfrequency range, at which point most high-permeability magneticmaterials (the type useful in the design of the CT-snubber) exhibit asignificant increase in impedance.

A study of the available commercial magnetic materials which might beuseful for the design of the CT-snubber has been conducted, and it wasdetermined that two separate approaches are available for the design ofthe CT-snubber. See especially, A. J. Hanson, J. A. Belk, S. Lim, C. R.Sullivan, and D. J. Perreault, “Measurements and Performance FactorComparisons of Magnetic Materials at High Frequency,” IEEE Trans. PowerElectron., vol. 31, no. 11, pp. 7909-7924, 2016. The first approach isto use a ferrite material with a high-permeability (in the range of2000-3000). As described by Snoek's Law, high-permeability soft ferritesbecome more dissipative with increased frequency than low-permeabilitysoft ferrites. In the case of the CT-snubber this increased dissipativebehavior translates to increased snubbing effectiveness. Therefore, whenusing a high-permeability magnetic material in the CT-snubber design,the magnetic material will provide the bulk of the snubbing influenceand the role of the RLC filter (212) is reduced to “tuning” oradjustment of the location of the suppression band. This design approachis useful for low-frequency applications because the magnetizinginductance of CT will be significant, improving the ability of theCT-snubber to measure long-duration pulses. The second approach is touse a magnetic material with lower permeability (in the range of100-500). Such materials generally are much less dissipative at highfrequencies than high-permeability soft ferrites, which translates toreduced snubbing effectiveness in the case of the CT-snubber. Therefore,when using a low-permeability magnetic material in the CT-snubberdesign, the magnetic material will provide little (if any) snubbinginfluence, and the RLC filter (212) is the primary means of suppressinghigh-frequency ringing. This design approach is especially useful forvery high-frequency applications, because the range of the suppressionband can be tightly controller. One disadvantage of this approach islimited ability to measure current at low frequencies, due to thelimited magnetizing inductance of the magnetic coupler.

Known snubbers, such as one taught by Kim et al. do not provide thesnubbing and simultaneous current measurement as in the presentlydisclosed principles. For example, Kim's approach lacks any type ofmagnetic core and uses instead winding in the printed circuit board(PCB) which is coupled through the air to the terminals of the primarycircuit by proximity. This approach has several disadvantages comparedto the presently disclosed principles, most of which accrue through thelack of a magnetic core. First, omission of the magnetic core means thatthe coupling between the primary and secondary circuits is sensitive tothe geometry of the circuit. This approach may require a much largersecondary coil than would be required with a magnetic core, and thecoupling effectiveness may be reduced if the relative positions of theprimary and secondary circuit are changed. Another more serious problemis that the lack of a magnetic core makes it impractical to effectivelymeasure the primary side current. Thus, Kim et. al do not claim anymeasurement capability for their proposed approach; only snubbingcapability. The lack of a magnetic core means that the magnetizinginductance of the coupling is so small that only very short pulses (lessthan one microsecond) could be reliably measured. In contrast, theCT-snubber has been demonstrated to provide high-frequency snubbingcapability simultaneously with measurement capability of pulses up to100 microseconds.

The measured impedance profile of the CT-snubber presented in FIG. 6(solid line) was imported into MATLAB and curve-fit to the impedanceprofile of an ideal parallel RLC network (dashed line). It should benoted that the experimental curves show a low-frequency phase near zero,while the model predicts a low-frequency phase at 90°. The reason forthis discrepancy is that the model represents an ideal parallel RLC,while the actual circuit contains an RLC network in series with a 2Ωburden resistor. Nevertheless, the RLC model obtained through curvefitting shows a reasonably good agreement with the measured curves up toabout 10 MHz, where the non-linear characteristics of the inductor corelikely begin to influence the measured impedance profile.

The impedance profile of FIG. 6 can be used to estimate the influencethat would be exerted on the main power circuit by the addition of theCT-snubber. This is the insertion impedance (Zin), or “loading” of themain power circuit due to the CT-snubber. As described previously, theCT-snubber is intended to minimize the insertion impedance in thefrequency range that represents the intended operation frequencies ofswitch-mode converters, and to maximize the insertion impedance in thehigh-frequency range that is occupied by undesirable, near-RF dynamicscaused by parasitic energy exchange, especially in systems withfast-switching WBG semiconductors. First, this analysis demonstratesthat an insertion impedance of approximately 20 mΩ would be reflectedinto the main power circuit at DC and low-frequency AC. This means thatthe nominal behavior of the converter should be largely unaffected bythe introduction of the CT-snubber, since most power electronicsapplications involve effective average power transfer in this frequencyrange (DC/DC converters and line-frequency inverters, for example).Second, impact to the main power circuit at a range of switchingfrequencies up to 100 kHz would be modest, with an insertion impedanceof 330 mΩ at the high end of this frequency range. Third, any content inthe 1-10 MHz range would be subject to the insertion impedance of the“suppression band”, which peaks at approximately 3.2Ω for thisparticular design. Spectral content in this range is usually unintendedand un-desired behavior for high-power SiC-based power electronicsconverters. Thus, the substantial impact expected in this frequencyrange due to the relatively high insertion impedance is expected tosubstantially improve the overall system dynamics of such a system,provided that the thermal behavior of the CT-snubber is sufficientlyrobust.

While certain values of R_(P), L_(P), and C_(P) are shown in FIG. 6, aCT-snubber design according to principles of the present disclosure isnot limited to these specific values. Simulation-based parametric sweepshave been performed to identify the range of RLC values which areexpected to be effective in the suppression of the high-frequencyspectral content commonly found in WBG-based systems. This analysisinvolved sweeping each of the three RLC filter parameter valuesindividually, while holding the remaining two parameter values constantat the nominal values presented herein. The resulting filterconfigurations were evaluated in simulation to determine whether thefrequency-dependent insertion impedance of each configuration would besuitable for the operation of the present invention. Specifically, it isdesired for Zin to remain below 0.5Ω at frequencies below 100 kHz; andfor Zin to be greater than 2Ω for the frequency range 1-10 MHz. Thisanalysis has shown that the following component ranges are expected tobe effective in the present invention: C_(P) values at 100 pF and below,with no lower limit (as shown in FIG. 7A), L_(P) values in the range of30 μH to 90 μH (as shown in FIG. 7B), and R_(P) values above 300Ω withno upper limit (as shown in FIG. 7C). An embodiment of the presentinvention with component values in these ranges is likely to be aseffective as the example design provided here. In addition, theinsertion impedance values suggested in this application are not limitedto the specific values indicated. As discussed above, for example, theinsertion impedance may be selected as 1.05 and 10 times themeasurement-band insertion impedance or may even vary from that rangedepending on the application.

FIG. 8 is a Clamped Inductive Load (CIL) test stand schematic thatincludes the CT-snubber implementation described herein.

The second portion of the empirical analysis performed for this effortinvolved introducing the CT-snubber into the high-power ClampedInductive Load (CIL) test stand designed specifically for transientevaluation of SiC multi-chip power modules and presented in FIG. 8. Anannotated picture of the hardware realization of this test stand ispresented in FIG. 9, which includes an indication of how the CT-snubberis integrated into the apparatus. This test stand incorporates ahigh-performance half-bridge SiC module rated at 1.2 kV and 350 A, asthe active semiconductor element. During the CIL test sequence, thelower switch (M2) is actively gated, and the upper switch (M1) is biasedoff such that the anti-parallel diodes in this position can serve as thefreewheeling path for the inductive current during the switch-offinterval. This test stand also contains provisions for accuratemeasurement of several important quantities associated with theactively-gated switch. For example, the drain-source voltage (VDS),gate-source voltage (VGS) of M2 are monitored with single-endedhigh-bandwidth voltage probes. The source current (IS) of M2 ismonitored by a high-bandwidth current-viewing-resistor (shunt)positioned adjacent to the location of the CT-snubber. This provides areference current measurement, which is used in this analysis toevaluate the current-sensing capabilities of the CT-snubber.

The large printed circuit board (PCB) shown in FIG. 9 represents anisolated energy storage bank, and the smaller “module board” containsthe interface to the device under test (DUT) as well as the necessarymetrology interfaces described previously. In this figure, themulti-chip module in use is not visible, as it is mounted to theunder-side of the module board. The two switch positions areindependently controlled by the two gate-drive modules shown. Theinstrumentation shunt shown in this picture is a carefully-calibrated100 mΩ resistor composed of ten 1Ω precision (1% tolerance) resistorswhich span the entire width of the negative bus plane. The output ofthis shunt is monitored by an oscilloscope via coaxial connection to thePCB. The coupling of the CT-snubber to the test stand is alsoillustrated in this figure.

This preliminary evaluation was performed with the CT-snubber coupled tothe power bus outside the module, as shown here, for proof of concept.The coupling mechanism is constructed from a pair of high-permeabilityferrite U-shaped-cores with a 10-turn secondary winding. The single-turnprimary winding consists of an aluminum standoff which also serves asthe mechanical mounting between the main board and the module board. Thesmall PCB which contains the burden resistor and filter portion of theCT-snubber is attached to the end of the wire leads extending from thecoupling mechanism. For evaluation of the CT-snubber concept, a set ofexperiments was conducted at a bus voltage of 600V and a load current of100 A.

The CIL experiment was first executed without the CT-snubber in place inorder to establish a baseline for the dynamics of the test stand. Theexpanded board-to-board spacing required for insertion of the CT-snubbercoupling mechanism was preserved for the baseline case; this has ameasurable effect on the inductance of the power loop bus andcontributes to undesirable ringing in the presence of fast-changingcurrent and voltage signals. The results of these experiments aredemonstrated in FIG. 10 and FIG. 11. FIG. 10 shows experimental turn-offwaveforms with and without the CT-snubber in place: (a) drain currentwaveforms, (b) drain-source voltage waveforms. In FIG. 10, significantringing at approximately 15 MHz is apparent for the baselineconfiguration (more than 25 cycles of oscillation are observable); whilethe CT-snubber configuration demonstrates significantly reduced ringing(fewer than 10 cycles of oscillation are observable). FIG. 11demonstrates the same type of transient comparison, but at thesemiconductor turn-on condition. FIG. 11 shows experimental turn-onwaveforms with and without the CT-snubber in place: (a) drain currentwaveforms, (b) drain-source voltage waveforms. As in the previous(turn-off) case, this comparison demonstrates a reduced number ofoscillatory cycles for the CT-snubber configuration compared to thebaseline case without the CT-snubber. Using the log-decrement method,the damping ratio of the voltage waveform can be calculated from theseempirical results and used to quantify the damping improvement achievedwith the CT-snubber. For example, at turn-off, the baselineconfiguration was determined to have a damping ratio of 0.034 (extremelyunder-damped). The turn-off voltage waveform for the CT-snubberconfiguration has a calculated damping ratio of 0.208, approximately sixtimes higher than the baseline case.

The final goal of this empirical study was to investigate the accuracyof the CT-snubber for the purpose of measuring the primary circuitcurrent. As identified previously, spectral components of the primarycurrent signal appearing in the measurement band are expected to becaptured with good fidelity by the CT-snubber as voltage transientsacross the burden resistor. In the experimental procedures outlinedabove, a high-voltage voltage probe was used to sample the instantaneousvoltage across the burden resistor (R_(B)) of the CT-snubber so that thecurrent projection of the CT-snubber could be compared to the referencecurrent measurement provided by a calibrated instrumentation-gradecurrent transducer. In this case, the CT-snubber inverse measurementgain was calculated to be 50 (I_(P)/V_(SENSE)=N_(S)/R_(B)). Afterapplying this scaling relationship to the voltage measured across theburden resistor, the resulting current projection from the CT-snubbercan be compared to the reference current measurement. FIG. 12 shows acomparison of module current using the reference current measurement(red trace) and the CT-snubber prediction (blue trace). This figuredemonstrates that the CT-snubber provides very good agreement with thereference current measurement.

Table 1, below, shows a synopsis of the previously describedexperimental results illustrating the effectiveness of a CT-Snubber forsuppression of high-frequency oscillation at semiconductor turn-off,according to principles of the present disclosure.

Configuration Ringing Frequency Damping Ratio Without Snubber 14.9 MHz0.034 With Snubber 11.1 MHz 0.208 6.1X Increase

As described herein, a technique for adding two complementary featuresto power electronics circuits based on multi-chip power modules:high-bandwidth current measurement and ringing suppression may beprovided. The design of the “CT-snubber” device, which can be viewed asan extension of the traditional current transformer, incorporates anadditional filter network which can be tuned to mitigate the undesirableparasitic-induced ringing of the type commonly observed in wide band-gapapplications during high-edge-rate switching transients. Preliminaryempirical results from a prototype CT-snubber designed as part of thiseffort indicate that this concept is viable both as a currentmeasurement apparatus, as well as a means for improving the transientresponse of power electronics applications. Further, it is believed thatthis type of circuit could be readily integrated into the housing ofmulti-chip modules, thereby simultaneously realizing improved dynamicperformance and in-situ current sensing suitable for applicationcontrol.

A CT-Snubber as described herein may allow for current sensing of“desired” spectral content and suppression of “undesired” spectralcontent; is amenable to implementation within multi-chip power modulesand is useful for SiC and other WBG-based systems; and can eliminate theneed to create a bus constriction for current measurement. The proposedCT-snubber may have a smaller form factor compared to the combination ofother circuits which are traditionally used to implement ringingsuppression and current measurement separately. As a result, theCT-snubber can be embedded into the MCPM geometry, taking advantage ofan existing bus constriction that is necessary at the interface betweenthe semiconductor device packaging and the remainder of the powerelectronics converter.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the invention. Thus, it isintended that the present invention cover the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

Throughout this application, various publications may have beenreferenced. The disclosures of these publications in their entiretiesare hereby incorporated by reference into this application in order tomore fully describe the state of the art to which this inventionpertains, including:

-   [1] A. N. Lemmon, A. Shahabi and K. Miskell, “Multi-branch    inductance extraction procedure for multi-chip power modules,” in    Proc. Workshop on Wide Bandgap Power Devices and Applications    (WiPDA), Fayetteville, A R, 2016, pp. 95-100.-   [2] S. Ziegler, R. C. Woodward, H. H. C. Iu and L. J. Borle,    “Current Sensing Techniques: A Review,” in IEEE Sensors Journal,    vol. 9, no. 4, pp. 354-376, April 2009.-   [3] S. Ziegler, “New current sensing solutions for low-cost    high-power-density digitally controlled power converters” Ph.D.    dissertation, school of Elect., Electronics and Comp. Engin., The    Univ. of Western Australia, Perth, Australia, 2009.-   [4] J. A. Ferreira, W. A. Cronje and W. A. Relihan, “Integration of    high frequency current shunts in power electronic circuits,” in IEEE    Transactions on Power Electronics, vol. 10, no. 1, pp. 32-37,    January 1995.-   [5] C. M. Johnson and P. R. Palmer, “Current measurement using    compensated coaxial shunts,” in IEEE Proceedings of Science,    Measurement and Technology, vol. 141, no. 6, pp. 471-480, November    1994.-   [6] Kwok-Wai Ma and Yim-Shu Lee, “Technique for sensing inductor and    DC output currents of PWM DC-DC converter,” in IEEE Transactions on    Power Electronics, vol. 9, no. 3, pp. 346-354, May 1994.-   [7] N. McNeill, N. K. Gupta and W. G. Armstrong, “Active current    transformer circuits for low distortion sensing in switched mode    power converters,” in IEEE Transactions on Power Electronics, vol.    19, no. 4, pp. 908-917, July 2004.-   [8] P. A. Tipler, An introduction to the Hall effect, Bell    Technologies, Inc., 2005. [Online]. Available:    http://www.fwbell.com.-   [9] R. S. Popovic, Z. Randjelovic, and D. Manic, “Integrated    Hall-effect magnetic sensors,” in Sensors and Actuators A: Physical,    vol. 91, pp. 46-50, 2001.-   [10] J. Kim, A Damping Scheme for switching Ringing of Full SiC    MOSFET by Air Core PCB circuit, in IEEE Transactions on Power    Electronics, Issue 99, 2017.-   [11] A. J. Hanson, J. A. Belk, S. Lim, C. R. Sullivan, and D. J.    Perreault, “Measurements and Performance Factor Comparisons of    Magnetic Materials at High Frequency,” IEEE Trans. Power Electron.,    vol. 31, no. 11, pp. 7909-7924, 2016.-   [12] A. Lemmon, R. Cuzner, J. Gafford, R. Hosseini, M. Mazzola,    “Methodology for Characterization of Common-Mode Conducted    Electromagnetic Emissions in Wide-Band-Gap Converters for Ungrounded    Shipboard Applications,” in IEEE Journal of Emerging and Selected    Topics in Power Electronics, vol. PP, no. 99, pp. 1-16, Jun. 2017.

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample only, and not limitation. It will be apparent to persons skilledin the relevant art that various changes in form and detail can be madetherein without departing from the spirit and scope of the presentinvention. Thus, the breadth and scope of the present invention shouldnot be limited by any of the above-described exemplary embodiments butshould be defined only in accordance with the following claims and theirequivalents.

What is claimed is:
 1. A current measurement and ringing suppressiondevice for use in a multi-chip power module comprising: a magnetic corecurrent transformer having an insertion impedance Zin and turns ratio N,a burden resistor having resistance R_(B) and a filter, the filtercomprising a capacitance C_(P), an impedance L_(P) and a resistanceR_(P); wherein C_(P), L_(P) and R_(P) are selected such that Zin remainsabove a minimum effective value across a known frequency range; whereincurrent measurement is observed from the voltage expressed across theburden resistor.
 2. The current measurement device of claim 1,comprising an insertion impedance Zin, where $\begin{matrix}{Z_{in} = {N^{2} \cdot ( {R_{B} + \frac{R_{P}}{1 + {j( {{\omega \; C_{P}R_{P}} - \frac{R_{P}}{\omega \; L_{P}}} )}}} )}} & (1)\end{matrix}$
 3. The current measurement device of claim 1, wherein Zinremains above 3Ω across the range of frequencies for which suppressionis desired.
 4. The current measurement device of claim 1, wherein C_(P)is lower than 100 pF.
 5. The current measurement device of claim 1,wherein L_(P) is between 30 μH and 90 μH.
 6. The current measurementdevice of claim 1, wherein R_(P) is greater than 300 n.
 7. A multichippower module comprising: a plurality of multi-chip power moduleterminals; a magnetic-core-based current transformer having an insertionimpedance Zin and turns ratio N, a burden resister having resistanceR_(B) and a filter, the filter comprising a capacitance C_(P), aninductance L_(P) and a resistance R_(P); wherein C_(P), L_(P) and R_(P)are selected such that Zin remains above a minimum effective valueacross a known frequency range; wherein current measurement is takenacross the burden resistor; wherein the magnetic core currenttransformer is positioned around a portion of at least one of themulti-chip power module terminals.
 8. The multichip power module ofclaim 7, wherein Zin remains above 3Ω across the range of frequenciesfor which suppression is desired.
 9. The multichip power module of claim7, wherein C_(P) is lower than 100 pF.
 10. The multichip power module ofclaim 7, wherein L_(P) is between 30 μH and 90 μH.
 12. The multichippower module of claim 7, wherein R_(P) is greater than 300Ω.
 13. Amethod of measuring current and snubbing in a multi-chip power module,the multi-chip power module comprising a magnetic core currenttransformer having an insertion impedance Zin and turns ratio N, aburden resister having resistance R_(B) and a filter, the filtercomprising a capacitance C_(P), an inductance L_(P) and a resistanceR_(P); wherein C_(P), L_(P) and R_(P) are selected such that|Zin|≈N ² ·R _(B) the method comprising: attaching the currenttransformer to the system for which measurement is desired; andmeasuring the voltage across the burden resistor R_(B); andmathematically scaling the voltage across the burden resistor R_(B) torepresent the predicted current value in the primary circuit.
 14. Themethod of claim 13, wherein Zin remains below 100 mΩ across the range offrequencies for which measurement is desired.
 15. The method of claim13, wherein C_(P) is lower than 100 pF.
 16. The method of claim 13,wherein L_(P) is between 30 μH and 90 μH.
 17. The method of claim 13,wherein R_(P) is greater than 300Ω.
 18. The method of claim 13, whereinR_(B) is less than 10 Ω.